Magnetic disk storage apparatus

ABSTRACT

Currents of sine waveforms can be fed through coils by a relatively small-sized circuit, and thereby, highly dense magnetic storage can be realized with less rotation variations and a driving control circuit of a motor rotating at a low noise level can be provided. A coil of one phase of a three-phase brushless motor is driven with full amplitude at which an applied voltage becomes equal to a source voltage, and a coil of one of other phases is driven with gradually changing voltages so that a current of sine waveform is delivered, and a coil of the remaining phase is driven by feedback control so that a total current flowing through all coils becomes a predetermined current value.

BACKGROUND OF THE INVENTION

The present invention relates to technology for driving-control of abrushless motor, and more particularly to technology effectively appliedto the formation of rotation drive current waveforms of the motor. Thepresent invention relates to technology effectively applied to a drivingcontrol apparatus of a spindle motor for rotationally driving disk typestorage media as in, e.g., a hard disk drive.

A hard disk drive is demanded to have the ability to read and writeinformation from and to magnetic disk as fast as possible, that is, theability to make access at high speed. To achieve this, it is importantto speed up disk rotation. Conventionally, a brushless DC multi-phasemotor called a spindle motor has been generally used to rotate magneticdisk in a hard disk drive. The magnetic disk is fast rotated by thespindle motor and a magnetic head for read and writing is brought nearto the surface of the rotating magnetic disk to write or readinformation while moving in a radius direction thereof.

In rotation driving control of a conventional spindle motor, a rotor hasbeen rotated by supplying coils of individual phases with square-wavepulse currents as shown in FIG. 15 that are out of phase with oneanother, by a driving circuit. FIG. 15 shows the waveform of current fedthrough one of three phases; currents having waveforms that are 120degrees out of phase with one another are fed through other two phases.Such a rotation driving method based on square-wave pulse currents hasthe advantage of easy current formation but also the disadvantage ofcausing rotation variations and noise due to torque ripple. It is knownthat a brushless motor can be rotated without causing rotationvariations and noise by using drive current waveforms of sine waveforms.Accordingly, an invention is proposed which smoothly rotates a rotor byfeeding pulse currents of sine waveforms through coils of individualphases (Japanese Published Unexamined Patent Application No. Hei9(1997)-37584).

SUMMARY OF THE INVENTION

However, in the above described technology, plural units of waveforminformation of one cycle of current waveforms to be formed are stored inROM (read only memory), depending on the load on the motor, and when auser selectively specifies one of them, the specified waveforminformation is read out to control coil drive currents, whereby currentsof desired sine waveforms are outputted. As a result, the amount ofhardware increases, and even if the load on the motor changes, since theduty of basic clock to form coil drive waveforms remains constant, phaseswitching of output currents cannot be smoothly performed in response toan increase or decrease in the output currents. This fact has beenrevealed by the present inventors.

An object of the present invention is to provide a magnetic disk unitthat can feed currents of sine waveforms through coils by a relativelysmall-sized circuit, and thereby, enables highly dense magnetic storagewith less rotation variations and has a spindle motor rotating at a lownoise level.

Another object of the present invention is to provide a magnetic diskunit that can smoothly change output currents in response to changes inthe load on a motor, and thereby, enables highly dense magnetic storagewith less rotation variations and has a spindle motor rotating at a lownoise level.

The above described objects and other objects and characteristics of thepresent invention will become apparent from the description of thisspecification and the accompanying drawings.

Typical ones of intentions disclosed by the present patent applicationwill be briefly described below.

A magnetic disk storage apparatus of this invention comprises: a firstmotor for rotating magnetic disk; a magnetic head for readinginformation from recording tracks on the magnetic disk; and a firstmotor driving control circuit for controlling drive currents of thefirst motor, wherein the first motor is a multi-phase brushless motor inwhich the potential of a center tap of the multi-phase brushless motoris made to be floating, and a driving control circuit of the first motorperforms driving by feedback control so that a coil of one of the phasesis driven with a full amplitude at which an applied voltage becomesequal to a source voltage, a coil of a second phase is driven withgradually changing voltages so that a current of sine waveform isdelivered, and a third coil is controlled so that a total currentflowing through all coils becomes a predetermined current value.

According to the above described means, motor coils can be drivenaccording to sine waveforms without causing power loss, whereby diskrotation variations are reduced, highly dense magnetic storage isenabled, and the motor can rotate at a low noise level.

Preferably, the first motor driving control circuit is provided with anarithmetic circuit that produces by predetermined operations a signaldriven with gradually changing voltages so that a current of sinewaveform is delivered. Accordingly, in comparison with the method ofholding all data corresponding to sine waveforms in memory, a circuitscale can be made smaller and the magnetic disk storage apparatus can beminiaturized.

Moreover, the first motor driving control circuit is constructed toproduce as a PWM signal a signal driven with gradually changing voltagesso that a current of sine waveform is delivered. A driving method basedon the PWM signal enables less power loss than a driving method based onlinearly changing currents.

The first motor driving control circuit is constructed to produce as aPWM signal a signal driven with the feedback control. Use of the PWMsignal can reduce power loss and enables still less rotation variationsbecause it can be driven with currents corresponding to changing loads.

Moreover, coil currents fed through coils of individual phases by thefirst motor driving control circuit are formed to have phases that arean predetermined electrical angle corresponding to coil inductance andinternal resistance ahead of the phases of back electromotive forcesinduced in the coils. Accordingly, the motor can be rotated with thegreatest driving torque.

Moreover, the first motor driving control circuit drives coils ofindividual phases so that phase switching timing is off zero-crosspoints of the back electromotive forces. Thereby, in the case wherephase switching control is performed by detecting zero-cross points ofback electromotive forces, the detection of incorrect zero-cross pointsdue to noise generated in the coils during phase switching can beprevented, so that highly accurate rotation control can be performed.

The first motor driving control circuit produces signals driven withgradually changing voltages by identical operations even if phasesdriven by the signals are different from each other so that currents ofsine waveforms are delivered. By producing drive control signals of allphases by identical operations, circuit configuration and arithmeticprograms can be simplified.

Moreover, in a magnetic disk storage apparatus comprising the firstmotor driving control circuit and a controller controlling the firstmotor driving control circuit, the first motor driving control circuitis constructed to perform control so that the total of currents fedthrough the coils of the phases matches a current indication valuesupplied from the controller, and a current indication value correctingcircuit is provided which corrects the current indication value, takinginto account fluctuations of the total current produced by the currentsfed through the coils of the phases being changed according to sinewaveforms. Accordingly, reaction of the control system to ripples ofcoil current resulting from driving the motor with a sine waveform canbe weakened, with the result that torque ripples can be reduced androtation variations can be further lessened.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 shows a driving circuit in a three-phase brushless motor to whichthe present invention is effectively applied, and an equivalent circuitof the motor;

FIG. 2 illustrates vector representation of applied voltage Vinput, coilvoltage Vcoil, and back electromotive force B-EMF;

FIG. 3 is a diagram showing a phase relationship among backelectromotive force B-EMF developed in coils Lm(U), Lm(V), and Lm(W) inthe equivalent circuit of FIG. 1, coil voltage Vcoil applied across thecoils, and applied voltage Vinput by the coil drive voltage sourcesVinput(U), Vinput(V), and Vinput(W);

FIG. 4 is a diagram showing an example of drive waveforms applied toindividual phases of a three-phase brushless motor by a motor drivingcontrol circuit to which the present invention is applied;

FIG. 5 is a timing diagram showing a mutual relationship of drivingmodes of coils of individual phases of a three-phase motor and aswitching timing;

FIG. 6 is a timing diagram enlarging a range from 90 to 270 degrees ofFIG. 5 to show a mutual relationship of driving modes of coils ofindividual phases, a switching timing, and duty changes of SP phase;

FIG. 7 is a block diagram showing one embodiment of a driving controlcircuit of a three-phase brushless motor to which the present inventionis applied;

FIG. 8 is a pattern diagram showing duty production patterns of SP phasein the motor driving control circuit of the embodiment of FIG. 7;

FIG. 9 is a flowchart showing an example of the procedure for producingthe duty of SP phase according to the patterns of FIG. 8;

FIG. 10 is a diagram for explaining changes of the duty of SP phaseproduced according to the procedure of FIG. 9;

FIG. 11 is a diagram showing waveforms of a PWM signal supplied to anoutput transistor driving a coil of SP phase produced according to theprocedure of FIG. 9;

FIG. 12 is a block diagram showing major parts of a second embodiment ofa three-phase brushless motor driving control circuit to which thepresent invention is applied;

FIG. 13 is a timing diagram showing a relationship between a currentindication value and current fluctuations developing when the motorcoils are driven with sine waveforms, in the second embodiment of thethree-phase brushless motor driving control circuit to which the presentinvention is applied;

FIG. 14 is a block diagram showing a configuration of a hard disk driveas one example of a system employing the motor driving control circuitto which the present invention is applied; and

FIG. 15 is a diagram showing an example of a drive waveform applied tocoils of individual phases by a driving control circuit of aconventional three-phase brushless motor.

DESCRIPTION OF THE PREFERRED EMBODIMENTS

Hereinafter, preferred embodiments of the present invention will bedescribed with reference to the accompanying drawings.

Before describing specific embodiments of the present invention, adriving principle of motor coils of the present invention will bedescribed using FIGS. 1 to 3. FIG. 1 shows a driving circuit in athree-phase brushless motor and an equivalent circuit of the motor. InFIG. 1, Lm(U), Lm(V), and Lm(W) respectively denote stator coils ofthree phases U, V, and W phases of a motor MT. Rm(U), Rm(V), and Rm(W)respectively denote internal resistances of phase coils Lm(U), Lm(V),and Lm(W). B-emf(U), B-emf(V), and B-emf(W) respectively denote backelectromotive forces of the phase coils Lm(U), Lm(V), and Lm(W). Ron(U),Ron(V), and Ron(W) respectively denote on resistances of outputtransistors making up a phase current output circuit that feeds currentsthrough the coils Lm(U), Lm(V), and Lm(W). Vinput(U), Vinput(V), andVinput(W) respectively denote drive voltage sources applied to thecoils.

FIG. 3 shows a phase relationship among waveforms of back electromotiveforces B-EMF developed in the coils Lm(U), Lm(V), and Lm(W) in theequivalent circuit of FIG. 1, coil voltage Vcoil applied across thecoils, and applied voltage Vinput by the coil drive voltage sourcesVinput(U), Vinput(V), and Vinput(W). When AC drive currents with thesame phase as the back electromotive forces B-EMF are fed through thecoils, the greatest torque is obtained.

However, even if drive voltages with the same phase as the backelectromotive forces B-EMF are applied to the coils, a phase lag occursin currents Icoil actually flowing through the coils because of internalresistance of the coils. Accordingly, as shown in FIG. 3, it isdesirable that coil voltage Vcoil of each phase is applied so that itsphase is Δθ coil ahead of that of the back electromotive forces B-EMFdeveloped in the coils Lm(U), Lm(V), and Lm(W), to match the phase ofcoil current Icoil to that of the back electromotive forces B-EMF. Sincevoltages Vinput applied by the drive voltage sources Vinput(U),Vinput(V), and Vinput(W) from outside the coils are also out of phasewith the coil voltages Vcoil of the individual phases, phase differencesmust be considered to decide the phases of the drive voltage sourcesVinput(U), Vinput(V), and Vinput(W).

A phase lead amount Δθcoil of the coil voltage Vcoil with respect to thephase of the back electromotive forces B-EMF is represented by thefollowing expression (1).Δθcoil=tan⁻¹(ω·Lm/Ron+Rm)=tan⁻¹{(2π·fB-EMF)·Lm/(Ron+Rm)}  (1)Δθcoil varies in value, depending on a motor used. In the expression(1), Lm denotes coil inductance and fB-EMF denotes the frequency of theback electromotive force B-EMF, that is, a required number ofrevolutions of a motor.

Next, assuming that a difference between the phase of the backelectromotive forces B-EMF of the coils and the phase of the drivevoltage sources Vinput(U), Vinput(V), and Vinput(W) is Δθ, the abovedescribed applied voltage Vinput is given as a synthetic vector of thecoil voltage Vcoil and the back electromotive forces B-EMF that arerepresented by vector, as shown in FIG. 2. Hence, if inductance Lm andinternal resistance Rm of the coils are determined from the motor used,a phase difference Δθcoil can be derived from the expression (1) and Δθcan be obtained from a vector diagram of FIG. 2. Accordingly, if drivewaveforms are formed by setting the phases of the drive voltage sourcesVinput(U), Vinput(V), and Vinput(W) to be Δθ ahead of that of the backelectromotive forces B-EMF, the greatest torque can be obtained. Thephase of the back electromotive forces B-EMF developed in the coils canbe obtained by detecting a zero-cross point of the back electromotiveforces.

In a motor driving circuit of an embodiment described below, an outputtransistor is controlled so that drive voltage waveforms of the phaserelationship as described above are applied to coils. Moreover, theoutput transistor is controlled by a PWM (pulse width modulation)system. In other words, a gate terminal of the output transistor iscontrolled by a PWM-controlled signal (pulse), whereby drive voltagewaveforms of the above described phase relationship are applied to thecoils.

As described previously, drive voltage waveforms applied to the coilsare desirably sine waveforms and their phases desirably have a timing asshown in FIG. 2. However, even if the coils are driven so as to satisfythe above condition, when the drive voltage waveforms shown in FIG. 3Care formed, if potential VCT of center tap CT, which is a commonconnection terminal of coils of three phases, is kept constant and sinewaveforms with the potential VCT as a center potential are formed andapplied to the coils, power loss will occur in a portion hatched in FIG.3C.

Accordingly, to reduce the power loss, we thought that a potential VCTof the center tap CT is set to be not fixed but floating so that a coildrive voltage around a portion in which a drive waveform of each phaseswings to its maximum amplitude is forcibly set to a source voltage Vccor ground potential GND (=0V). FIG. 4A shows a waveform produced when acoil drive voltage around a portion in which a drive waveform of eachphase swings to its maximum amplitude is forcibly set to the sourcevoltage GND (=0V). FIG. 4B shows a waveform produced when a coil drivevoltage around a portion in which a drive waveform of each phase swingsto its maximum amplitude is forcibly set to the source voltage Vcc.

It will be understood from FIG. 4 that the case (A) of FIG. 4 eliminatespower loss at a lower hatched portion in FIG. 2C and the case of (B)eliminates power loss at an upper hatched portion in FIG. 2C.Accordingly, by using the drive waveforms of FIG. 4A or FIG. 4B, higherpower efficiency can be obtained than the case where a potential VCT ofthe center tap CT is fixed to drive the coils with sine waveforms asshown in FIG. 3C. In FIGS. 4A and 4B, it is because a potential VCT ofthe center tap CT floats that waveforms at portions not set to Vcc orGND appear to be not sine waveforms. Use of the potential VCT of thecenter tap CT as reference, that is, differences between-the potentialVCT of the center tap CT and the potential of individual waveforms tellthat the drive waveforms change according to sine waveforms.

In this embodiment, the above described driving system is furtheradvanced to the system of using waveforms as shown in FIG. 4C fordriving. Employing this system contributes to simplification of hardwareconfiguration. Waveforms of FIG. 4C are formed by combining waveformscut from portions from 0 to 37.5 degrees, 97.5 to 157.5 degrees, 217.5to 277.5 degrees, and 337.5 to 360 degrees from FIG. 4A and portions of37.5 to 97.5 degrees, 157.5 to 217.5 degrees, and 277.5 to 337.5 degreesfrom FIG. 4B.

Cutting is not performed in units of 60 degrees such as 0 to 60 degrees,60 to 120 degrees, 120 to 180 degrees, and 180 to 240 degrees, and soforth. This is because, as seen from FIG. 1, 60, 120, 180, 240, and 300degrees are respectively zero-cross points of back electromotive force,and in this embodiment, as described later, since zero-cross points ofback electromotive force of the coils are detected to perform drivingcontrol, switching of currents of individual phases at such positionscauses noise to occur in back electromotive force, disabling correctdetection of zero-cress positions.

Next, a description will be made of a specific method of forming thedrive waveforms as shown in FIG. 4. In FIG. 4C, the symbols “SP”, “PWM”,and “F” provided in the vicinity of a waveform of each phase indicatethe type of a method of forming each waveform. A different symbolindicates a different formation method. Hereinafter, a method of formingeach waveform will be described in order.

First, a waveform marked with the symbol “F” is formed by forciblydriving an output transistor into a full amplitude level. Specifically,the output transistor driving a coil of a phase corresponding to awaveform marked with the symbol “F” is applied with a control signal ofhigh level to its gate terminal continuously for a required time(corresponding to the length of the F waveform), thereby applying Vcc(e.g., 12V) or GND (0V) to a driving terminal of the coil.

Next, a waveform marked with the symbol “SP” is produced by operationsin an arithmetic circuit and formed by the output transistor beingdriven by a PWM-controlled signal. As shown in FIG. 4, waveforms markedwith “SP” exist two for each of a right upward direction and a rightdownward direction in a range of an electrical angle 60 degrees cut asdescribed previously, and are of identical shape or vertically symmetricshape. Therefore, they are produced by only two arithmetic expressions.If only waveforms of a right upward direction and a right downwarddirection are formed by operations, other waveforms or part of waveformscan be formed by feedback control based on current detection or fullamplitude driving of the output transistor.

Thereby, in comparison with the conventional system of forming waveformsthroughout 360 degrees according to ROM data, the system of thisembodiment forms waveforms more easily and reduces the amount ofhardware. A specific example of an operation method by an arithmeticcircuit will be described in detail later; for waveforms marked with thesymbol “SP”, PWM signals are formed which turn the output register on oroff by 16 or 32 pulses within a range of an electrical angle 60 degrees.Specifically, pulse width is controlled to become gradually wider forright upward portions and gradually narrower for right downwardportions.

FIG. 5B shows by which of “F”, “PWM”, and “SP” methods outputtransistors driving coils of three phases U, V, and W in which backelectromotive force B-EMF changes as indicated by (A) form waveforms ata proper timing. In the drawing, “upper arm” denotes a transistor of thepower voltage Vcc side of an output transistor of a corresponding phase,and “lower arm” denotes an output transistor of the GND side. A box thatis described across the boundary between “upper arm” and “lower arm” andmarked with the symbol “D” denotes that both an output transistor of theVcc side and an output transistor of the GND side are turned off. Thereason that periods are thus provided in which both an output transistorof the Vcc side and an output transistor of the GND side are turned offis to eliminate influence of drive voltages applied to coils whenzero-cross points of back electromotive forces are detected, in orderthat only the back electromotive forces are observed to detect thezero-cross points.

FIG. 6A is an enlarged view of duty changes of pulses of a signal fordriving an output transistor of PWM phase subjected to feedback controlbased on current detection in a portion of 90 to 270 degrees of FIG. 5showing waveforms in a range from 0 to 360 degrees and duty changes ofpulses of a signal for driving an output transistor of SP phasecontrolled by operations in an arithmetic circuit. The duty controlapplies to not all cases and phases are automatically adjusted as shownby the arrow A according to the magnitude of output current. The phaseadjustment is made based on coil current values detected in the range ofthe first preceding 60 degrees. FIG. 6B is an enlarged view of a portionfrom 90 to 270 degrees of FIG. 5B showing the timings of waveformforming methods in the range from 0 to 360 degrees.

Next, a waveform marked with “PWM” is formed based on a currentdetection and current comparison function of a motor driving controlcircuit of the embodiment. Specifically, the motor driving controlcircuit of the embodiment is provided with a current detection resistorRNF provided so that the sum of currents flowing through three coils Lu,Lv, and Lw flows to detect a total of them, and a current detectiondifferential amplifier that detects a potential difference across thecurrent detection resistor RNF to detect the magnitude of current. Tocontrol an output current, a PWM signal is produced which detects adifference between a coil current value detected by the currentdetection differential amplifier and a current indication value suppliedfrom a controller (CPU) (not shown) and drives the output transistors sothat the difference is 0.

For example, when a detected current is smaller than the currentindication value, the duty of the PWM signal is increased to allow morecurrent to flow through the coils, while, when the detected current islarger than the current indication value, the duty of the PWM signal isreduced to decrease current flowing through the coils. By repeating thisoperation, a waveform marked with the symbol “PWM” is formed. Dutycontrol of the PWM signal is performed based on the magnitude of outputcurrent detected in the preceding cycle. Thereby, the phase of dutycontrol of the PWM signal, that is, the phase of sawtooth waveform ofFIG. 6A is automatically adjusted according to an output currentdetected in the preceding cycle.

Furthermore, in this embodiment, waveforms in the range of electricalangle 60 degrees are formed by, e.g., 16 PWM pulses. In other words, theoutput transistors are turned on and off 16 times by 16 pulses formedwhen a rotor rotates by an electrical angle of 60 degrees, and therespective widths of the 16 pulses are changed according to the detectedcurrent value, whereby waveforms marked with the symbol “PWM” areformed. Since such drive pulse feedback control based on currentdetection has been performed by a motor driving control circuit of theconventional PWM control system as well, a drive waveform applied to anyone coil of three phases by the same circuits and procedure asconventional ones.

FIG. 7 shows an embodiment of the present invention effectively appliedto a driving control circuit of a spindle motor used in a hard diskstorage apparatus. The whole circuit shown in FIG. 7 is formed on onesemiconductor substrate such as a single-crystal silicon, except coilsLu, Lv, and Lw of the motor.

In FIG. 7, the reference numeral 11 designates a current output circuitsuccessively feeding current to the coils Lu, Lv, and Lw of athree-phase brushless motor; 12, an output current control circuit thatproduces a PWM signal for controlling an output current and supplies itto the current output circuit 11; RNF, a current detection resistorconnected to the current output circuit 11 so that the sum of currentsflowing through the three coils Lu, Lv, and Lw flows to detect a totalof them; 13, a current detection differential amplifier that detects apotential difference across the current detection resistor RNF to detectthe magnitude of current; and 14, an AD conversion circuit that performsAD conversion for an output voltage of the current detectiondifferential amplifier to produce a digital signal.

Reference numeral 15 designates a back electromotive force detectingcircuit that detects back electromotive forces of the coils Lu, Lv, andLw developing in output terminals u, v, and w of the current outputcircuit 11, and center tap CT to output a signal indicating a zero-crosspoint; 16, a phase difference detecting circuit that detects a phasedifference between a signal indicating a zero-cross point of backelectromotive force outputted from the back electromotive forcedetecting circuit 15 and a signal indicating a zero-point of an outputcurrent outputted from the output current control circuit 12; 17, a loopfilter that performs phase compensation of a main line; and 18, anoscillation circuit that oscillates at a frequency (about 100 kHz)corresponding to a value (digital code) of the loop filter 17. An outputof the oscillation circuit 18 is used as a reference clock for producingthe PWM signal in the output current control circuit 12.

PLL (phase locked loop) is formed by a feedback route established by thephase difference detecting circuit 16, loop filter 17, oscillationcircuit 18, output current control circuit 12, and phase differencedetecting circuit 16 back from the output current control circuit 12.The PLL controls oscillation operation of the oscillation circuit 18 sothat the phase of a signal indicating a zero-cross point of backelectromotive force outputted from the back electromotive forcedetecting circuit 15 matches the phase of a signal outputted from theoutput current control circuit 12, thereby locking the frequencies ofvoltage waveforms (1 to 2 kHz) applied to the coils.

Reference numeral 19 designates an AD conversion circuit that performsAD conversion for a back electromotive force outputted from the backelectromotive force detecting circuit; 20, a conduction start controlcircuit that decides a conduction start phase, based on a backelectromotive force induced in a nonconduction phase and detected by theback electromotive force detecting circuit 15 when a short pulse towhich the rotor does not respond is fed from one phase to another by thecurrent output circuit 11, based on an output of the AD conversioncircuit 19 when the motor is standing; 21, a serial port that sends andreceives data to and from a microcomputer (CPU) (not shown).

The serial port 21 receives a serial clock SCLK supplied from the CPU, acurrent indication value of a spindle motor, and information about anoperation mode, and produces control signals inside the driving controlcircuit, based on received mode information.

Reference numeral 22 designates a sequencer that controls the whole ofcircuits shown in FIG. 7; 23, an arithmetic circuit that produces a dutycontrol signal for forming drive waveforms of SP phase; 24, a currentdifference detecting circuit that detects a difference between a coilcurrent value detected by the current detection differential amplifier13 and a current indication value supplied via the serial port 21 fromthe CPU; and 25, a filter that produces a value corresponding to acurrent difference detected based on an output of the current differencedetecting circuit 24 while making phase compensation. Current differenceinformation outputted from the filter 25 and waveform informationproduced in the arithmetic circuit 23 are supplied to the output currentcontrol circuit 12, where a PWM signal is produced to drive the outputtransistors and supplied to the current output circuit 11 to controloutput currents to be fed to the coils.

Output current Iout is represented byIout={(Vcc×Duty)−Bemf}/RLwhere Duty is the duty (ratio of pulse width to one cycle) of PWMsignal, Bemf is coil back electromotive force, and RL is coilresistance. Accordingly, changes of PWM signal cause coil output currentIout to be controlled according to the above expression.

Next, a more specific method of producing waveforms (hereinafterreferred to as waveforms of SP phase) marked with the symbol SP in FIG.4C will be described using FIG. 8.

First, a description is made of the case where coil back electromotiveforce B-EMF and coil voltage Vcoil are not out of phase with each other.Suppose that an output current and a current indication value from theCPU match and the duty of a control signal for producing waveforms(hereinafter referred to as waveforms of PWM phase) marked with thesymbol PWM in FIG. 4C is constant (e.g., 70%). FIGS. 8A and 8B show arelationship between back electromotive force B-EMF in that case and theduty of a control signal for producing waveforms of SP phase. FIG. 8Ashows the duty of PWM phase at the left scale and the duty of SP phasein a direction opposite to PWM phase at the right scale, representingchanges in PWM phase duty (constant) and SP phase duty, correspondinglyto the respective scales.

In FIG. 8, in an electrical angle of 90 degrees, the back electromotiveforce of U phase is zero, just in the middle of the back electromotiveforces of V phase and W phase. At this time, V phase is PWM phase, Uphase is SP phase, and V phase is a phase (hereinafter referred to as Fphase) driven into full amplitude. Accordingly, the duty of U phase,which is SP phase, is 65% of a complement D1 (=100−D0/2) for 100% ofjust the half (D0/2) of the duty (D0 (=70%) of V phase, which is PWMphase. Since waveforms of U phase change to full amplitude in a sectionfrom 65% to 100%, the duty may be changed from 65% to 100%. In thisembodiment, since duty changes at this time could be linearly made withonly small errors, linear changes are alternatively employed to simplifycontrol.

If a waveform of U phase reaches duty 100% at 120 degrees, thereafter, Uphase is switched to the F phase of full amplitude driving, W phase,which has been hitherto F phase, is switched to SP phase, and the dutyof the control signal is linearly changed from 100% to 65%. Phaseswitching is made again at an electrical angle of 150 degrees such thatV phase, which has been hitherto PWM phase, is switched to SP phase, theduty of the control signal is linearly changed from 65% to 100%, Wphase, which has been SP phase, is switched to F phase, and U phase,which has been F phase, is switched to PWM phase; this is continued upto an electrical angel of 180 degrees. At an electrical angle of 180degrees, F phase, which has been hitherto F phase, is switched to SPphase, the duty of the control signal is linearly changed from 100% to65%, and V phase, which has been SP phase, is switched to F phase. Atthis time, U phase is left to be PWM phase.

The above waveforms are true for the case where coil back electromotiveforce B-EMF and coil voltage Vcoil are not out of phase with each other.If coil back electromotive force B-EMF and coil voltage Vcoil are out ofphase with each other, the waveforms are formed as shown in FIG. 8C.That is, with the same tilt as the tilt of duty change of each SP phasein FIG. 8B, a starting point is advanced by phase Δθ to control the dutyof each SP phase. By this arrangement, the phase of a coil voltage Vcoilof each phase leads the phase of back electromotive force B-EMF and coilcurrent Icoil is driven in phase with back electromotive force B-EMF, sothat the greatest torque can be produced.

In the case of FIG. 8B, phase switching takes place at electrical anglesof 30, 90, 150, 210, 270, and 330 degrees, and these points correspondto zero-cross points of back electromotive force B-EMF. For this reason,phase switching at these points may cause noise to occur in backelectromotive force and disable correct detection of zero-cross points.Accordingly, as shown in FIG. 8D, it is desirable to control duties soas to delay phase switching timing by Δoffset (e.g., an electrical angle7.5 degrees).

Since a V-shaped waveform of FIG. 8D highly resembles a waveform of SPphase in a range from 100 to 160 degrees in FIG. 4C, it is understoodthat a waveform similar to a desired waveform (sine waveform as viewedfrom the center tap) can be produced by the above described duty controlmethod. It can be easily determined that waveforms of SP phase in otherportions, which are vertically symmetrical, can be realized by reversingthe positive/negative relationship of the above described duty control.In the motor driving control circuit of the embodiment of FIG. 7, theabove described duty control is achieved by the arithmetic circuit 23and a PWM control circuit within the output current control circuit 12in coordination.

Next, the procedure of operations in the arithmetic circuit 23 forproducing waveforms of the above SP phase is described using a flowchartof FIG. 9. The meanings of variables used in the procedure by theflowchart are shown in FIG. 10. As seen from FIG. 10, PWM controlfollowing the flowchart is not continuous but is performed in 16 stagesaccording to 16 PWM pulses in one conduction period (electrical angle of60 degrees). The number of PWM pulses in one conduction period isarbitrary.

The PWM control circuit, which produces a predetermined number (e.g.,16) of PWM pulses in a conduction period of each phase and applies themto an output transistor, successively adds on time (e.g., high levelperiod) of the 16 PWM pulses in one conduction period to find total ontime Ton-total, and calculates an average value PWMave by dividing thetotal time (Ton-total) by the number of pulses DIV at phase switching(step S1). An output value of the AD conversion circuit 14 in oneconduction period is successively added and the total of them is dividedby the number of pulses DIV at phase switching to find an average valueItotalave of a total output current (step S2).

Next, a coefficient CIADJ (=Δθ/Itotal) inputted from the CPU via aserial port and the average output current Itotalave calculated in stepS2 are multiplied to obtain a phase lead amount Δθ1 of an appliedvoltage Vinput applied to a coil (step S3). Δθ and Itotal, instead ofthe coefficient CIADJ, may be given from the CPU to obtain a coefficientby operations in the motor driving control circuit.

In the next step S4, a value Δθ2 (=Δθ1−Δoffset) is calculated bysubtracting a delay amount Δoffset of phase switching timing fromzero-cross point from the phase lead amount Δθ1 obtained in step S3. Theaverage value PWMave of total on time of PWM pulses calculated in stepS1 is divided by the number of pulses DIV to obtain an average dutychange amount Δndown (=PWMave/DIV) per PWM pulse. In the next step S6,the average duty change amount Δndown obtained in step S5 is multipliedby Δθ2 obtained in step S4 to find an decrease amount ΔCNT from theaverage value PWMave of total on time of PWM pulses.

The average value PWMave of total on time of PWM pulses is halved toobtain a loopback point duty (D1 of FIG. 8B) of the SP phase in the casewhere it is assumed that there is no phase lag, and the value issubtracted by the decrease amount ΔCNT obtained in step S6 to calculatethe duty SSN0 of a PWM pulse applied to a first SP phase after phaseswitching (step S7). Thereafter, the duty of PWM pulse a second time orlater, namely, on time SSNd is decided by subtracting the change amountΔndown obtained in step S5 from on time SSNd−1 of a previous PWM pulse(step S8).

In the next step S9, it is judged whether on time SSNd decided in stepS8 is equal to or smaller than 0, and step S8 is repeated until SSNd isequal to or smaller than 0, whereby the duties of SP phase in a downperiod indicated by the symbol Tdown in FIG. 10 are successivelyoutputted. If on time SSNd is equal to or smaller than 0, control istransferred to step S10, where a predetermined change amount Δndown1 isadded to on time SSNu−1 of the previous PWM pulse to decide the duty ofthe next PWM pulse, namely, on time SSNu.

In the next step S11, it is decided whether the number N of producedpulses reaches the number DIV of pulses in one conduction period, andstep S10 is repeated until N and DIV match, whereby the duties of SPphase in an up period indicated by the symbol Tup in FIG. 10 aresuccessively outputted. A line indicated by a dashed line A is a dutychange line of the SP phase in the case where it is assumed that thereis no phase lag, and corresponds to the waveform in FIG. 8B. The dutiesSSNd and SSNu calculated in the steps S8 and S10 are all outputted asone's complement numbers (1-SSNd) or (1-SSNu). This is done to convertduties calculated at the left scale of FIG. 8 to the right scale.

FIG. 11A shows PWM pulses in the down period Tdown of the SP phase ofFIG. 10 produced based on the duty SSNd calculated in step S8 of FIG. 9,and FIG. 11B shows PWM pulses in the up period Tup of the SP phase ofFIG. 10.

The PWM pulses of FIG. 11A are applied to the gate terminal of an outputtransistor (N-MOS) of the Vcc side driving the coil (Lv) forming SPphase in a period from 37.5 to 55 degrees of FIG. 4C. If the transistoris P-MOS, inversion signals of the PWM pulses of FIG. 11A are applied tothe gate terminal. The PWM pulses of FIG. 11A are applied to the gateterminal of an output transistor (N-MOS) of the GND side driving thecoil (Lu) forming SP phase in a period from 97.5 to 115 degrees of FIG.4C.

The PWM pulses of FIG. 11B are applied to the gate terminal of an outputtransistor (N-MOS) of the Vcc side driving the coil (Lu) forming SPphase in a period from 55 to 97.5 degrees of FIG. 4C. If the transistoris P-MOS, inversion signals of the PWM pulses of FIG. 11B are applied tothe gate terminal. The PWM pulses of FIG. 11B are applied to the gateterminal of an output transistor (N-MOS) of the GND side driving thecoil (Lw) forming SP phase in a period from 115 to 157.5 degrees of FIG.4C. By the above described method, waveforms of SP phase that arevertically symmetrical can be formed according to an identicalprocedure, using identical values.

FIG. 12 shows a configuration of major parts of a motor driving controlcircuit in a second embodiment of the present invention.

As described previously, the motor driving control circuit of the firstembodiment is provided with the current detection resistor RNE fordetecting a total current flowing through the three coils Lu, Lv, and Lwand the differential amplifier 13, wherein a difference between adetected coil current value and a current indication value supplied fromthe controller (CPU) outside the drawing is detected, and a PWM signalis produced to drive the output transistor so as to make the differencezero so that output current fed through the coils is subjected tofeedback control. On the other hand, in the motor driving controlcircuit of the present embodiment, since coils of three phases of themotor are driven with three sine waveforms that are 120 degrees out ofphase with one another, a total current Itotal flowing through the motorfluctuates and forms a rippled waveform indicated by a solid line B inFIG. 13.

If the total current is detected by the current detection resistor RNFand the differential amplifier 13 and compared with a current indicationvalue SPNCRNT (constant within a short time) given from the CPU, judgingthat an error occurs, the feedback control system of the output currentcontrol circuit 12 reacts to the ripple and changes output current.Since there is a delay in the current control system, torque ripplebecomes worse.

Accordingly, in the embodiment of FIG. 12, an error current detectingcircuit 24 is provided with a correction arithmetic circuit 26 thatcorrects a current indication value by multiplying a current indicationvalue SPNCRNT given from the CPU by a coefficient, and a selector 27.The coefficient multiplied by the current indication value SPNCRNT is avalue such as, e.g., 1.1, according to average regulation of outputcurrent. The selector 27 selects between a value with a currentindication value SPNCRNT multiplied by a coefficient and a value withthe current indication value SPNCRNT not multiplied by a coefficient andsupplies the value to an add circuit 28 that finds a difference betweenthe value and an output of the AD conversion circuit.

Switching timing of the selector 27 can be automatically obtained fromphase switching timing of the output current control circuit 12.Specifically, taking delay in the control system into account, theselector 27 may be subjected to switching control so that the selector27 selects a value with a current indication value SPNCRNT multiplied bya coefficient as in FIG. 13A in accordance with the timing when the ADconversion circuit 14 outputs current values corresponding to ridgedportions of the total current Itotal indicated in FIG. 13B, and selectsa value with the current indication value SPNCRNT not multiplied by acoefficient in accordance with the timing when the AD conversion circuit14 outputs current values corresponding to valley portions of the totalcurrent Itotal. The coefficient multiplied by the current indicationvalue SPNCRNT may be a value smaller than 1 such as e.g., 0.9, to switchthe selector in the reverse timing of the above.

As in this embodiment, by changing the current indication value SPNCRNTaccording to the fluctuation of coil total current Itotal, reaction ofthe control system to ripples of coil current can be weakened, with theresult that torque ripples resulting from driving the motor with a sinewaveform can be reduced. Although, in this embodiment, a currentindication value SPNCRNT is changed at two levels, plural correctionarithmetic circuits 26 that corrects a current indication value bymultiplying a current indication value SPNCRNT by a coefficient may beprovided and appropriately selected by the selector 27 according to thefluctuation of total current Itotal so that the current indication valueSPNCRNT is changed at three levels or more.

FIG. 14 is a block diagram showing a configuration of a hard disk driveas one example of a magnetic disk system including a spindle motorcontrol system employing a motor driving control circuit to which thepresent invention is applied, and a magnetic head driving controlsystem.

In FIG. 14, 210 designates a spindle motor driving control circuit,which is configured as shown in FIG. 7, drives and controls the spindlemotor 310, and rotates magnetic disk at a predetermined speed. Thespindle motor driving control circuit 210 operates according to controlsignals such as a current indication value SPNCRNT supplied from acontroller 260 comprising a microcomputer and performs servo control forthe spindle motor 310 so as to keep relative speed of a magnetic headconstant.

Reference numeral 320 designates an arm having a magnetic head(including a write magnetic head and a read magnetic head) HD and 330designates a carriage rotatably holding the arm 320. The voice coilmotor 340 moves the carriage 330 to move the magnetic head, and a VCMdriving circuit 100 performs servo control for the voice coil motor 340to align the center of the magnetic head with the center of track.

Reference numeral 220 designates a read/write IC that amplifies currentcorresponding to a magnetic change to send a read signal to a signalprocessing circuit (data channel processor) 230 or amplifies a writepulse signal from the signal processing circuit 230 to output drivecurrent of the magnetic head HD. Reference numeral 240 designates a harddisk controller that gets read date sent from the signal processingcircuit 230 to perform error correcting processing, and performserror-correcting encoding processing for write data from a host tooutput the result to the signal processing circuit 230. The abovedescribed signal processing circuit 230 performs modulation/demodulationprocessing suitable for digital magnetic recording and signal processingincluding waveform shaping with magnetic recording characteristics inmind, and reads position information from a read signal of the magnetichead HD.

Reference numeral 250 designates an interface controller that performsdata exchange and control between this system and external apparatuses,and the hard disk controller 240 is connected to a host computer such asa microcomputer of a personal computer body via the interface controller250. Reference numeral 270 designates a cache memory for temporarilystoring read data read at high speed from magnetic disk. A systemcontroller 260 comprising a microcomputer judges an operation mode froma signal supplied from the hard disk controller 240, controls variousparts of the system according to the operation mode, and calculates asector position and the like from address information supplied from thehard disk controller 240.

As described above, the present invention made by the inventor has beendescribed in detail based on preferred embodiments. It goes withoutsaying that the present invention is not limited to the above describedpreferred embodiments, but may be modified in various ways withoutdeparting from the spirit and the scope of the present invention. Forexample, in the motor driving circuit of the above describedembodiments, although the sensorless method is employed to detect arotor stop position and decide a conduction start phase by detectingback electromotive force, a rotor stop position may be detected using ahole sensor or the like. The motor may be not a three-phase motor butmultiple-phase motor.

Although, in the embodiments, waveforms of SP phase are produced byoperations in an arithmetic circuit, a memory to store datacorresponding to waveforms may be provided so that waveforms areproduced by successively reading the data from the memory. Moreover,although, in the embodiments, a MOS transistor is used as an outputtransistor, a bipolar transistor can be used as an output transistor.Moreover, although, in the embodiments, the full-wave driving method isdescribed, the present invention can apply to the half-wave drivingmethod also.

Although the present invention has been described as to application to amotor driver apparatus of a hard disk storage apparatus, which is anapplication field of the present invention, the present invention is notlimited to such a field and can be widely used in a motor drivingcontrol apparatus driving brushless motors such as, e.g., a motor forrotating a polygon mirror of a laser beam printer and an axial fanmotor.

Effects obtained by typical ones of inventions disclosed by the presentpatent application are briefly described below.

According to the present invention, currents of sine waveforms can befed through coils by a relatively small-sized circuit. With thisconstruction, highly dense magnetic storage can be realized with lessrotation variations and a magnetic disk unit provided with a spindlemotor rotating at a low noise level can be achieved.

1-9. (cancelled)
 10. A magnetic disk storage apparatus comprising: a motor for rotating a magnetic disk; a magnetic head disposed for reading information from recording tracks on the magnetic disk; and a motor driving control circuit that controls drive currents of the motor, wherein the motor is a multi-phase brushless motor, and wherein the motor driving control circuit of the motor performs driving by feedback control so that for a coil of one phase a first drive is implemented with an amplitude at which an applied voltage becomes equal to a source voltage, for a coil of a second phase a second drive is implemented with gradually and linearly changing voltage so that a current of sine waveform is delivered, and for a coil of a third phase a third drive is implemented so that a total current flowing through all coils becomes a predetermined current value by a PWM drive.
 11. The magnetic disk storage apparatus according to claim 10, wherein the motor driving control circuit includes an arithmetic circuit that produces, by predetermined operations, a signal driven with gradually and linearly changing voltages so that a current of sine waveform is delivered.
 12. The magnetic disk storage apparatus according to claim 11, wherein the motor driving control circuit produces, as a PWM signal, a signal driven with gradually and linearly changing voltage so that a current of sine waveform is delivered.
 13. The magnetic disk storage apparatus according to claim 11, wherein the motor driving control circuit produces, as a PWM signal, a signal driven with the feedback control.
 14. The magnetic disk storage apparatus according to claim 10, wherein coil currents fed through coils of individual phases by the motor driving control circuit are formed to have phases that are of predetermined electrical angle, corresponding to coil inductance and internal resistance, ahead of phases of back electromotive forces induced in the coils.
 15. The magnetic disk storage apparatus according to claim 14, wherein the motor driving control circuit drives coils of individual phases so that phase switching timing is off zero-crossing points of the back electromotive forces.
 16. The magnetic disk storage apparatus according to claim 10, wherein the motor driving control circuit produces signals driven with gradually and linearly changing voltages by identical operation even if phases driven by the signals are different from each other so that currents of sine waveforms are delivered.
 17. The magnetic disk storage apparatus according to claim 10, including a controller that controls the motor driving control circuit, said controller being constructed to perform control so that the total of currents fed through the coils of the phases matches a current indication value supplied from the controller, and including a current indication value correcting circuit that corrects the current indication value, taking into account fluctuations of the total current produced by the currents fed through the coils of the phases being changed according to sine waveforms.
 18. The magnetic disk storage apparatus according to claim 10, wherein the motor is a three-phase brushless motor.
 19. The magnetic disk storage apparatus according to claim 10, wherein in a first stage, the first drive is implemented for the coil of the one phase, the second drive is implemented for the coil of the second phase, and the third drive is implemented for the coil of the third phase, wherein in a stage 60 degrees ahead of the first stage, the third drive is implemented for the coil of the one phase, the first drive is implemented for the coil of the second phase, and the second drive is implemented for the coil of the third phase, and wherein in a stage 120 degrees ahead of the first stage, the second drive is implemented for the coil of the one phase, the third drive is implemented for the coil of the second phase, and the first drive is implemented for the coil of the third phase. 